As is known in the art, switching mixers are frequently used in RF design, and in order to make the switches turn on and off quickly and cleanly these mixers are frequently driven with a large local oscillator (LO) signal such that the signal is compressed. More particularly, the LO signal is a square wave or a very large sinusoidal signal, such that the input voltage swing is larger than the linear region of the transistors. Once the linear region of the transistor is exceeded, the voltage swing will be compressed and appear more like a square wave as the odd-order harmonics increase in power.
A significant problem with the compressed LO is the odd harmonic content which will mix with the desired signal and create stray signals at the output of the mixer. The compression of the LO signal can either occur prior to the mixer (for example, in a buffer amplifier), or the compression can occur within the mixer as the input is over driven, or a combination of the two. FIG. 1A shows how copies of the baseband signal will be located at odd multiples of the LO at the output of the mixer. These copies can be reduced by filtering, or as proposed in this document, the 3rd and 5th harmonics can be reduced by harmonic reject mixing.
The harmonic reject mixer replaces the single mixer with a combination of three mixers, each fed with a copy of the LO signal that has been shifted in phase. This approximates the behavior of a single mixer replacing the single compressed LO signal with a 3-bit quantized sinusoid signal, as shown in FIG. 1B. The quantized sinusoid signal has its first harmonic at the 7th multiple of the fundamental, which greatly eases the filtering requirements for reducing the stray signals at the mixer output [see J. Weldon, et al., “A 1.75-GHz Highly Integrated Narrow-Band CMOS Transmitter with Harmonic-Reject Mixers,” IEEE Journal of Solid-State Circuits. Vol. 36, No. 12, pp. 2003-2015, 2001]. The filtering may even be able to be eliminated because the power of the 7th multiple of the fundamental is much weaker than the 3rd or 5th multiples.
As mentioned above, the compressed LO signals are desirable because they allow the current-commutating nature of the switching mixers to quickly steer current from one switch to the other while minimizing conversion loss. The quantized sinusoidal signal can be created from compressed waves by summing them together with specific phase shifts (0, 45, 90) and amplitude adjustments, as seen in FIG. 2. The rejection of the 3rd and 5th harmonics of the compressed waves can be seen by inspecting the Fourier series representation of the scaled and shifted square waves. Summing these three series together, with f2(t) multiplied by sqrt(2) will result in the cancellation of the 3rd and 5th harmonic terms. In the case of the harmonic reject mixer, the summation of the three signals occurs at the output of the three mixers.
Another way to understand the poly-phase mixing, the phase of the third harmonic of the LO changes 3 times faster than that of the fundamental LO [see S. Lerstaveesin, et al., “A 48-860 MHz CMOS Low-IF Direct-Conversion DTV Tuner,” IEEE Journal of Solid-State Circuits. Vol. 43, No. 9, pp. 2013-2024. 2008]. The fundamental has the phase offsets of 0, 45, and 90 degrees, so the third harmonic will have phase offsets of 0, 135 and 270 degrees. Notice in the middle diagram of FIG. 3 that the LO vectors of the 3rd harmonic sum destructively. The same destructive summing also happens for the 5th harmonic.
The harmonic reject mixer is implemented by feeding each of three parallel double balanced switching mixers with one of the three phase shifted square waves, and summing the output, as shown in FIG. 4. The transconductance stage of each mixer is driven by the baseband signal, and the summing at the output is where the harmonic cancelation occurs. The middle mixer's bias current was increased by a factor of sqrt(2) to increase its gain and provide the sqrt(2) amplitude scaling necessary for the harmonic cancelation. The output signal (IF) is effectively the baseband signal multiplied by the 3-bit amplitude quantized sinusoid.
As is also known in the art, a phase rotator is based on the concept of a vector modulator, depicted in FIGS. 5A and 5B. The four vectors used are the differential pairs of a cosine and sine, also known as quadrature (0, 90, 180, and 270 in phase). Each differential pair is fed to an independent variable gain transconductance amplifier, with the capability to multiply by negative values by swapping the differential inputs to the gain stage. After the input quadrature voltages are amplified and converted into current, the currents are summed and converted back into a voltage in either a configuration of passive devices or an output buffer. The resulting amplitude and phase of the output signal is determined by the gain and sign assigned to the quadrature inputs, producing a signal with amplitude R and phase θ as depicted in FIG. 5B. [see H. Wang and A. Hajimiri, “A Wideband CMOS Linear Digital Phase Rotator,” IEEE 2007 Custom Integrated Circuits Conference (CICC). Vol. TP-30, pp. 1-4, 2007]
FIG. 6 shows a more complete representation of how a single phase rotator setting relates to a nominal reference state, resulting in a relative amplitude and phase shift. For discussion, it will here be assumed that the reference state for the phase rotator is the state defined by the I-path set at maximum positive gain with the Q-path set to zero gain. If one were to plot this reference point on the diagram in FIG. 6, it would be a point residing at (1, 0). To illustrate a phase rotator state other than the reference, FIG. 6 shows a single configuration with a relative amplitude equal to the radius of the blue circle, and a phase shift equivalent to the degrees of rotation from the reference point at (1,0) to the point.
A typical configuration for a phase rotator is shown in FIG. 7. For compatibility with differential circuits, a circuit to produce quadrature signals from a single differential pair is often used. In the example, a quadrature all-pass filter performs this function, but a poly phase circuit or other active or passive techniques can be used in its place [see K. Koh and G. Rebeiz, “0.13-um CMOS Phase Shifters for X-, Ku-, and K-Band Phased Arrays,” IEEE Journal of Solid State Circuits, Vol. 42, No. 11, pp. 2535-2546, 2007]. Inside the analog differential adder, the input signal is optionally shifted by 180 degrees by routing the differential pair such that the two signals swap paths. These signals are then amplified by two separate variable-gain amplifiers (one of the in-phase or “I” path, another for the quadrature or “Q” path). To produce the output, the signals are summed together as described previously and in this case the output is matched to a 50 ohm characteristic impedance with a configuration of passive lumped elements. In this particular example, the variable gain amplifiers are comprised of digitally-selectable gain states and a logic encoder to map digital input signals to specific states that correspond to desired phase shifts. The harmonic reject mixer requires that the three phase shifted square waves have a very specific phase delay (0, 45, 90) and amplitude scaling (1, sqrt(2), 1) for ideal cancellation of the 3rd and 5th harmonics.
Amplitude and phase mismatch in the LO path can come from a number of areas, including: layout differences, passive component tolerances, and LO generation method [see N. Moseley, et al., “A Two-Stage Approach to Harmonic Rejection Mixing Using Blind Interference Cancellation,” IEEE Transactions on Circuits and Systems-II: Express Briefs. Vol. 55, No. 10, pp. 966-970. 2008]. Most designs try to minimize the mismatch with good layout practices or by increasing the size of the devices so that mismatch constitutes a smaller percent of the device's total area. These methods are sensitive to process variation, require more area and more DC and LO power to drive the devices. These methods are also usually limited to around 40 dBc of harmonic rejection [see S. Lerstaveesin, et al., “A 48-860 MHz CMOS Low-IF Direct-Conversion DTV Tuner,” IEEE Journal of Solid-State Circuits. Vol. 43, No. 9, pp. 2013-2024. 2008]. Other methods of phase/gain LO mismatch compensation have been proposed that use either digital or analog compensation in the signal path [see N. Moseley, et al., “A Two-Stage Approach to Harmonic Rejection Mixing Using Blind Interference Cancellation,” IEEE Transactions on Circuits and Systems-II: Express Briefs. Vol. 55, No. 10, pp. 966-970. 2008 and H. Cha, et al., “A CMOS Harmonic Rejection Mixer with Mismatch Calibration Circuitry for Digital TV Tuner Applications,” IEEE Microwave and Wireless Components Letters. Vol. 18, No. 9, pp. 617-619. 2008]. The analog compensation method uses adjustable resistors in and between the signal paths before the summing at the output of the mixer. These programmable resistors adjust small amounts of gain mismatch to improve the harmonic rejection. The digital method uses sensors and a digital algorithm to generate a cancellation signal through a digital to analog converter, and combines the error signal with the mixer output to minimize the interference.
The inventors have recognized that better harmonic rejection in the mixer can be achieved by using a phase rotator to provide small amounts of phase and gain adjustment in the LO path. The phase rotator can produce large shifts in phase in addition to fine tuning, which makes it useful to provide the necessary 45 and 90 degree phase shifts as well. Finally, the phase rotator can account for I/Q mismatch if this “Harmonic Reject Mixer with Active Mismatch Compensation in the Local Oscillator Path” is used in an image reject transmitter or receiver topology, as show in FIG. 10 (transmitter). With this method and system, two quadrature LO signals (LO1 and LO2 for double up conversion) may be distributed to the harmonic mixers of stage 2 and mixers of stage 1. A set of three phase rotators provide the necessary coarse phase shift (0, 45, 90), fine phase and amplitude mismatch adjustments, and fine I/Q phase mismatch adjustments for the harmonic mixers in stage 2. A single phase rotator may provide the fine I/Q phase mismatch adjustment for the mixer in stage 1. This combination of mixers and gain/phase adjustment greatly improve harmonic and image rejection of this transmitter.
This method and system of compensating a harmonic reject mixer is digitally tunable due to the phase rotators digital control, and may be combined with an algorithm and/or analog sensors to automatically manage the image rejection. The compact layout of the phase-rotator allows it to be used three times with each harmonic reject mixer without significantly increasing the die area. The phase rotator can also be digitally compensated to maintain a given phase shift over a range of LO frequencies, which allows the harmonic reject mixer and gain/phase mismatch compensation to operate in wideband transmitter or receiver architectures.
The combination of a harmonic reject mixer and a phase rotator in the LO path enables good harmonic rejection to be maintained in an environment of non-ideal LO generation techniques, or mismatches in the LO path resulting from manufacturing process, temperature, and tolerance variation. This method and system also allows harmonic rejection to be maintained over varying local oscillator frequencies, which makes this mixer suitable for wide-band transmitter or receiver architectures. If this mixer and mismatch compensation were used in an image reject transmitter or receiver topology, the same mismatch compensation method may be used to eliminate I/Q mismatch, and improve the image rejection of that transmitter or receiver. Previous methods rely on increasing device sizes and power consumption, precise layout, or compensation in the signal path.
In accordance with one embodiment of the disclosure, a harmonic rejection mixer is provided having: a plurality of phase rotators fed by a common local oscillator signal, such local oscillator signal having a reference frequency, each one of the phase rotator output signals having a common frequency related to the reference frequency and having different relative phase shifts; and a plurality of mixer sections, each one of the mixer sections being fed an input and a corresponding one of the plurality of output signals to mix the input signal with the corresponding one of the plurality of output signals fed thereto.
In one embodiment, a combiner is provided for combining the mixer signal from the plurality of mixer sections into a composite output signal.
In one embodiment, a detector is provided for detecting energy in a harmonic of the composite signal and for adjusting the output signal of the phase rotator to reduce the selected harmonic of the composite signal.
In one embodiment, a harmonic rejection mixer is provided having: a plurality of phase rotators fed by a common input local oscillator signal having a reference frequency, each one of the phase rotators separating the common local oscillator signal into a pair of channels with an in-phase signal in one of the channels having a ninety degree phase shift relative to a quadrature signal in the other one of the channels, each one of the channels providing a selected gain to the in-phase signal and quadrature signal therein and wherein the phase rotator combines the gain provided in-phase signal and quadrature signal to produce a composite local oscillator signal having a selected one of a plurality of phase shifts relative to the common input local oscillator signal; and a plurality of mixer sections, each one of the mixer sections being fed the composite local oscillator signal of a corresponding one of the plurality of phase rotators and a common input signal to translate the frequency of the input signal by a frequency related to frequency of the local oscillator signal.
The details of one or more embodiments of the disclosure are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the disclosure will be apparent from the description and drawings, and from the claims.
Like reference symbols in the various drawings indicate like elements.